Load control device for a light-emitting diode light source

ABSTRACT

A load control device for controlling the intensity of a light source (e.g., a light-emitting diode light source) reduces visible flickering in the light source when a target intensity of the light source is dynamically changing. The load control device includes a load regulation circuit operable to conduct a load current through the electrical load and to control the amount of power delivered to the electrical load. The load control device also includes a controller for adjusting the average magnitude of the load current to a target current. The controller pulse-width modulates the magnitude of the load current between a first current less than the target current and a second current greater than the target current. When the target current is dynamically changing, the controller is operable to adjust to the average magnitude of the load current towards the sum of the target current and a supplemental signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a non-provisional application of commonly-assignedU.S. Provisional Application No. 61/668,759, filed Jul. 6, 2012,entitled LOAD CONTROL DEVICE FOR A LIGHT-EMITTING DIODE LIGHT SOURCE,the entire disclosure of which is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a load control device for alight-emitting diode (LED) light source, and more particularly, to anLED driver for controlling the intensity of an LED light source.

2. Description of the Related Art

Light-emitting diode (LED) light sources (i.e., LED light engines) areoften used in place of or as replacements for conventional incandescent,fluorescent, or halogen lamps, and the like. LED light sources maycomprise a plurality of light-emitting diodes mounted on a singlestructure and provided in a suitable housing. LED light sources aretypically more efficient and provide longer operational lives ascompared to incandescent, fluorescent, and halogen lamps. In order toilluminate properly, an LED driver control device (i.e., an LED driver)must be coupled between an alternating-current (AC) source and the LEDlight source for regulating the power supplied to the LED light source.The LED driver may regulate either the voltage provided to the LED lightsource to a particular value, the current supplied to the LED lightsource to a specific peak current value, or may regulate both thecurrent and voltage.

LED light sources are typically rated to be driven via one of twodifferent control techniques: a current load control technique or avoltage load control technique. An LED light source that is rated forthe current load control technique is also characterized by a ratedcurrent (e.g., approximately 350 milliamps) to which the peak magnitudeof the current through the LED light source should be regulated toensure that the LED light source is illuminated to the appropriateintensity and color. In contrast, an LED light source that is rated forthe voltage load control technique is characterized by a rated voltage(e.g., approximately 15 volts) to which the voltage across the LED lightsource should be regulated to ensure proper operation of the LED lightsource. Typically, each string of LEDs in an LED light source rated forthe voltage load control technique includes a current balance regulationelement to ensure that each of the parallel legs has the same impedanceso that the same current is drawn in each parallel string.

It is known that the light output of an LED light source can be dimmed.Different methods of dimming LEDs include a pulse-width modulation (PWM)technique and a constant current reduction (CCR) technique. Pulse-widthmodulation dimming can be used for LED light sources that are controlledin either a current or voltage load control mode. In pulse-widthmodulation dimming, a pulsed signal with a varying duty cycle issupplied to the LED light source. If an LED light source is beingcontrolled using the current load control technique, the peak currentsupplied to the LED light source is kept constant during an on time ofthe duty cycle of the pulsed signal. However, as the duty cycle of thepulsed signal varies, the average current supplied to the LED lightsource also varies, thereby varying the intensity of the light output ofthe LED light source. If the LED light source is being controlled usingthe voltage load control technique, the voltage supplied to the LEDlight source is kept constant during the on time of the duty cycle ofthe pulsed signal in order to achieve the desired target voltage level,and the duty cycle of the load voltage is varied in order to adjust theintensity of the light output. Constant current reduction dimming istypically only used when an LED light source is being controlled usingthe current load control technique. In constant current reductiondimming, current is continuously provided to the LED light source,however, the DC magnitude of the current provided to the LED lightsource is varied to thus adjust the intensity of the light output.Examples of LED drivers are described in greater detail incommonly-assigned U.S. patent application Ser. No. 12/813,908, filedJun. 11, 2010, and U.S. patent application Ser. No. 13/416,741, filedMar. 9, 2012, both entitled LOAD CONTROL DEVICE FOR A LIGHT-EMITTINGDIODE LIGHT SOURCE, the entire disclosures of which are herebyincorporated by reference.

In addition, some LED light sources comprise forward converters fordriving the LED light sources to control the load current conductedthrough the LED light source. Forward converters comprise a transformerhaving a primary winding coupled to at least one semiconductor switchand a secondary winding operatively coupled to the LED light source. Thesemiconductor switch is rendered conductive and non-conductive toconduct a primary current through the primary winding and to thustransfer power to the secondary winding of the transformer. Forwardconverters typically comprise an optocoupler for coupling a feedbacksignal on the secondary side of the transformer to the primary side ofthe transformer, such that a controller can control the semiconductorswitch is response to the feedback signal. However, there is a need fora forward converter that is able to control the magnitude of the loadcurrent through an LED light source without the need for an optocoupler.

SUMMARY OF THE INVENTION

As described herein, a load control device for controlling the intensityof a light source (e.g., a light-emitting diode light source) reducesvisible flickering in the light source when a target intensity of thelight source is dynamically changing. The load control device includes aload regulation circuit operable to conduct a load current through theelectrical load and to control the amount of power delivered to theelectrical load. The load control device also includes a controller.When a target current is in a steady state, the controller is operableto pulse-width modulate the magnitude of the load current between acurrent less than the target current and a current greater than thetarget current to adjust an average magnitude of the load current to thetarget current. When the target current is dynamically changing, thecontroller is operable to pulse-width modulate the magnitude of the loadcurrent between a sum of a supplemental signal and a current less thanthe target current and a sum of the supplemental signal and a currentgreater than the target current to adjust the average magnitude of theload current towards the sum of the target current and the supplementalsignal.

In addition, a method for controlling the amount of power delivered toan electrical load is also described herein. The method comprises: (1)conducting a load current through the electrical load; (2) when a targetcurrent is in a steady state, adjusting the average magnitude of theload current to the target current by pulse-width modulating themagnitude of the load current between a current less than the targetcurrent and a current greater than the target current; and (3) when thetarget current is dynamically changing, adjusting the average magnitudeof the load current towards a sum of the target current and asupplemental signal by pulse-width modulating the magnitude of the loadcurrent between a sum of the supplemental signal and a current less thanthe target current and a sum of the supplemental signal and a currentgreater than the target current.

Other features and advantages of the present invention will becomeapparent from the following description of the invention that refers tothe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified block diagram of a light-emitting diode (LED)driver for controlling the intensity of an LED light source.

FIG. 2 is a simplified schematic diagram of an isolated forwardconverter and a current sense circuit of an LED driver.

FIG. 3 is an example diagram illustrating a magnetic core set of anenergy-storage inductor of a forward converter.

FIG. 4 shows example waveforms illustrating the operation of a forwardconverter and a current sense circuit when the intensity of an LED lightsource is near a high-end intensity.

FIG. 5 shows example waveforms illustrating the operation of a forwardconverter and a current sense circuit when the intensity of an LED lightsource is near a low-end intensity.

FIG. 6 is an example plot of a relationship between an offset time and atarget intensity of an LED driver.

FIG. 7 is a simplified flowchart of a control procedure executedperiodically by a controller of an LED driver.

FIG. 8 is an example plot of a relationship between the offset time andthe target intensity of an LED driver.

FIG. 9 shows an example waveform of a load current conducted through anLED light source when a target current of an LED driver is at asteady-state value.

FIG. 10 shows an example waveform of the load current conducted throughthe LED light source when the target current of the LED driver is beingincreased with respect to time.

FIG. 11 shows an example waveform of a periodic ramp signal that isadded to a target current of an LED driver.

FIG. 12 is a flowchart of a load current adjustment procedure executedperiodically by a controller of an LED driver.

DETAILED DESCRIPTION OF THE INVENTION

The foregoing summary, as well as the following detailed description ofthe preferred embodiments, is better understood when read in conjunctionwith the appended drawings. For the purposes of illustrating theinvention, there is shown in the drawings an embodiment that ispresently preferred, in which like numerals represent similar partsthroughout the several views of the drawings, it being understood,however, that the invention is not limited to the specific methods andinstrumentalities disclosed.

FIG. 1 is a simplified block diagram of a load control device, e.g., alight-emitting diode (LED) driver 100, for controlling the amount ofpower delivered to an electrical load, such as, an LED light source 102(e.g., an LED light engine), and thus the intensity of the LED lightsource. The LED light source 102 is shown as a plurality of LEDsconnected in series but may comprise a single LED or a plurality of LEDsconnected in parallel or a suitable combination thereof, depending onthe particular lighting system. In addition, the LED light source 102may alternatively comprise one or more organic light-emitting diodes(OLEDs). The LED driver 100 comprises a hot terminal H and a neutralterminal that are adapted to be coupled to an alternating-current (AC)power source (not shown).

The LED driver 100 comprises a radio-frequency (RFI) filter circuit 110for minimizing the noise provided on the AC mains and a rectifiercircuit 120 for generating a rectified voltage V_(RECT). The LED driver100 further comprises a boost converter 130, which receives therectified voltage V_(RECT) and generates a boosted direct-current (DC)bus voltage V_(BUS) across a bus capacitor C_(BUS). The boost converter130 may alternatively comprise any suitable power converter circuit forgenerating an appropriate bus voltage, such as, for example, a flybackconverter, a single-ended primary-inductor converter (SEPIC), a Ćukconverter, or other suitable power converter circuit. The boostconverter 120 may also operate as a power factor correction (PFC)circuit to adjust the power factor of the LED driver 100 towards a powerfactor of one. The LED driver 100 also comprises a load regulationcircuit, e.g., an isolated, half-bridge forward converter 140, whichreceives the bus voltage V_(BUS) and controls the amount of powerdelivered to the LED light source 102 so as to control the intensity ofthe LED light source between a low-end (i.e., minimum) intensity L_(LE)(e.g., approximately 1-5%) and a high-end (i.e., maximum) intensityL_(HE) (e.g., approximately 100%). Alternatively, the load regulationcircuit could comprise any suitable LED drive circuit for adjusting theintensity of the LED light source 102.

The LED driver 100 further comprises a control circuit, e.g., acontroller 150, for controlling the operation of the boost converter 130and the forward converter 140. The controller 150 may comprise, forexample, a digital controller or any other suitable processing device,such as, for example, a microcontroller, a programmable logic device(PLD), a microprocessor, an application specific integrated circuit(ASIC), or a field-programmable gate array (FPGA). The controller 150generates a bus voltage control signal V_(BUS-CNTL), which is providedto the boost converter 130 for adjusting the magnitude of the busvoltage V_(BUS). The controller 150 receives from the boost converter130 a bus voltage feedback control signals V_(BUS-FB), which isrepresentative of the magnitude of the bus voltage V_(BUS).

The controller 150 also generates drive control signals V_(DRIVE1),V_(DRIVE2), which are provided to the forward converter 140 foradjusting the magnitude of a load voltage V_(LOAD) generated across theLED light source 102 and the magnitude of a load current I_(LOAD)conducted through the LED light source to thus control the intensity ofthe LED light source to a target intensity L_(TRGT). The LED driver 100further comprises a current sense circuit 160, which is responsive to asense voltage V_(SENSE) that is generated by the forward converter 140and is representative of the magnitude of the load current I_(LOAD). Thecurrent sense circuit 160 is responsive to a signal-chopper controlsignal V_(CHOP) (which is received from the controller 150), andgenerates a load current feedback signal V_(I-LOAD) (which is a DCvoltage representative of the magnitude of the load current I_(LOAD)).The controller 150 receives the load current feedback signal V_(I-LOAD)from the current sense circuit 160 and controls the drive controlsignals V_(DRIVE1), V_(DRIVE2) to adjust the magnitude of the loadcurrent I_(LOAD) to a target load current I_(TRGT) to thus control theintensity of the LED light source 102 to the target intensity L_(TRGT).The target load current I_(TRGT) may be adjusted between a minimum loadcurrent I_(MIN) and a maximum load current I_(MAX).

The controller 150 is coupled to a memory 170 for storing theoperational characteristics of the LED driver 100 (e.g., the targetintensity L_(TRGT), the low-end intensity L_(LE), the high-end intensityL_(HE), etc.). The LED driver 100 may also comprise a communicationcircuit 180, which may be coupled to, for example, a wired communicationlink or a wireless communication link, such as a radio-frequency (RF)communication link or an infrared (IR) communication link. Thecontroller 150 may be operable to update the target intensity L_(TRGT)of the LED light source 102 or the operational characteristics stored inthe memory 170 in response to digital messages received via thecommunication circuit 180. Alternatively, the LED driver 100 could beoperable to receive a phase-control signal from a dimmer switch fordetermining the target intensity L_(TRGT) for the LED light source 102.The LED driver 100 further comprises a power supply 190, which receivesthe rectified voltage V_(RECT) and generates a direct-current (DC)supply voltage V_(CC) for powering the circuitry of the LED driver.

FIG. 2 is a simplified schematic diagram of a forward converter 240 anda current sense circuit 260, e.g., the forward converter 140 and thecurrent sense circuit 160 of the LED driver 100 shown in FIG. 1. Theforward converter 240 comprises a half-bridge inverter circuit havingtwo field effect transistors (FETs) Q210, Q212 for generating ahigh-frequency inverter voltage V_(INV) from the bus voltage V_(BUS).The FETs Q210, Q212 are rendered conductive and non-conductive inresponse to the drive control signals V_(DRIVE1), V_(DRIVE2), which arereceived from a controller (e.g., the controller 150). The drive controlsignals V_(DRIVE1), V_(DRIVE2) are coupled to the gates of therespective FETs Q210, Q212 via a gate drive circuit 214 (e.g., partnumber L6382DTR, manufactured by ST Microelectronics). The controllergenerates the inverter voltage V_(INV) at a constant operating frequencyf_(OP) (e.g., approximately 60-65 kHz) and thus a constant operatingperiod T_(OP). However, the operating frequency f_(OP) may be adjustedunder certain operating conditions. The controller adjusts the dutycycle DC of the inverter voltage V_(INV) to adjust the magnitude of theload current I_(LOAD) and thus the intensity of an LED light source 202.The forward converter 240 may be characterized by a turn-on timeT_(TURN-ON) from when the drive control signals V_(DRIVE1), V_(DRIVE2)are driven high until the respective FET Q210, Q212 is renderedconductive, and a turn-off time T_(TURN-OFF) from when the drive controlsignals V_(DRIVE1), V_(DRIVE2) are driven low until the respective FETQ210, Q212 is rendered non-conductive.

The inverter voltage V_(INV) is coupled to the primary winding of atransformer 220 through a DC-blocking capacitor C216 (e.g., having acapacitance of approximately 0.047 μF), such that a primary voltageV_(PRI) is generated across the primary winding. The transformer 220 ischaracterized by a turns ratio n_(TURNS) (i.e., N₁/N₂) of approximately115:29. The sense voltage V_(SENSE) is generated across a sense resistorR222, which is coupled series with the primary winding of thetransformer 220. The FETs Q210, Q212 and the primary winding of thetransformer 220 are characterized by parasitic capacitances C_(P1),C_(P2), C_(P3).

The secondary winding of the transformer 220 generates a secondaryvoltage, which is coupled to the AC terminals of a full-wave dioderectifier bridge 224 for rectifying the secondary voltage generatedacross the secondary winding. The positive DC terminal of the rectifierbridge 224 is coupled to the LED light source 202 through an outputenergy-storage inductor L226 (e.g., having an inductance ofapproximately 10 mH), such that the load voltage V_(LOAD) is generatedacross an output capacitor C228 (e.g., having a capacitance ofapproximately 3 μF).

FIG. 3 is an example diagram illustrating a magnetic core set 290 of anenergy-storage inductor (e.g., the output energy-storage inductor L226of the forward converter 240 shown in FIG. 2). The magnetic core set 290comprises two E-cores 292A, 292B, and may comprise part numberPC40EE16-Z, manufactured by TDK Corporation. The E-cores 292A haverespective outer legs 294A, 294B and inner legs 296A, 296B. Each innerleg 296A, 296B may have a width w_(LEG) (e.g., approximately 4 mm). Theinner leg 296A of the first E-core 292A has a partial gap 298A (i.e.,the magnetic core set 290 is partially-gapped), such that the inner legs296A, 296B are spaced apart by a gap distance d_(GAP) (e.g.,approximately 0.5 mm). The partial gap 298A may extend for a gap widthw_(GAP), e.g., approximately 2.8 mm, such that the gap extends forapproximately 70% of the leg width w_(LEG) of the inner leg 296A.Alternatively, both of the inner legs 296A, 296B could comprise partialgaps. The partially-gapped magnetic core set 290 shown in FIG. 3 allowsthe output energy-storage inductor L226 of the forward converter 240shown in FIG. 2 to maintain continuous current at low load conditions(e.g., near the low-end intensity L_(LE)).

FIG. 4 shows example waveforms illustrating the operation of a forwardconverter and a current sense circuit, e.g., the forward converter 240and the current sense circuit 260 shown in FIG. 2. The controller drivesthe respective drive control signals V_(DRIVE1), V_(DRIVE2) high toapproximately the supply voltage V_(CC) to render the respective FETsQ210, Q212 conductive for an on time T_(ON) at different times (i.e.,the FETs Q210, Q212 are not conductive at the same time). When thehigh-side FET Q210 is conductive, the primary winding of the transformer220 conducts a primary current I_(PRI) to circuit common through thecapacitor C216 and sense resistor R222. Immediately after the high-sideFET Q210 is rendered conductive (at time t₁ in FIG. 4), the primarycurrent I_(PRI) conducts a short high-magnitude pulse of current due tothe parasitic capacitance C_(P3) of the transformer 220 as shown in FIG.4. While the high-side FET Q210 is conductive, the capacitor C216charges, such that a voltage having a magnitude of approximately half ofthe magnitude of the bus voltage V_(BUS) is developed across thecapacitor. Accordingly, the magnitude of the primary voltage V_(PRI)across the primary winding of the transformer 220 is approximately equalto approximately half of the magnitude of the bus voltage V_(BUS). Whenthe low-side FET Q212 is conductive, the primary winding of thetransformer 220 conducts the primary current I_(PRI) in an oppositedirection and the capacitor C216 is coupled across the primary winding,such that the primary voltage V_(PRI) has a negative polarity with amagnitude equal to approximately half of the magnitude of the busvoltage V_(BUS).

When either of the high-side and low-side FETs Q210, Q212 areconductive, the magnitude of an output inductor current I_(L) conductedby the output inductor L226 and the magnitude of the load voltageV_(LOAD) across the LED light source 202 both increase with respect totime. The magnitude of the primary current I_(PRI) also increases withrespect to time while the FETs Q210, Q212 are conductive (after theinitial current spike). When the FETs Q210, Q212 are non-conductive, theoutput inductor current I_(L) and the load voltage V_(LOAD) bothdecrease in magnitude with respective to time. The output inductorcurrent I_(L) is characterized by a peak magnitude I_(L-PK) and anaverage magnitude I_(L-AVG) as shown in FIG. 4. The controller increasesand decreases the on times T_(ON) of the drive control signalsV_(DRIVE1), V_(DRIVE2) (and the duty cycle DC of the inverter voltageV_(INV)) to respectively increase and decrease the average magnitudeI_(L-AVG) of the output inductor current I_(L) and thus respectivelyincrease and decrease the intensity of the LED light source 102.

When the FETs Q210, Q212 are rendered non-conductive, the magnitude ofthe primary current I_(PRI) drops toward zero amps (e.g., as shown attime t₂ in FIG. 4 when the high-side FET Q210 is renderednon-conductive). However, current may continue to flow through theprimary winding of the transformer 220 due to the magnetizing inductanceL_(MAG) of the transformer (which conducts a magnetizing currentI_(MAG)). In addition, when the target intensity L_(TRGT) of the LEDlight source 102 is near the low-end intensity L_(LE), the magnitude ofthe primary current I_(PRI) oscillates after either of the FETs Q210,Q212 is rendered non-conductive due to the parasitic capacitancesC_(P1), C_(P2) of the FETs, the parasitic capacitance C_(P3) of theprimary winding of the transformer 220, and any other parasiticcapacitances of the circuit, such as, parasitic capacitances of theprinted circuit board on which the forward converter 240 is mounted.

The real component of the primary current I_(PRI) is representative ofthe magnitude of the secondary current I_(SEC) and thus the intensity ofthe LED light source 202. However, the magnetizing current I_(MAG)(i.e., the reactive component of the primary current I_(PRI)) also flowsthrough the sense resistor R222. The magnetizing current I_(MAG) changesfrom negative to positive polarity when the high-side FET Q210 isconductive, changes from positive to negative polarity when the low-sideFET Q212 is conductive, and remains constant when the magnitude of theprimary voltage V_(PRI) is zero volts as shown in FIG. 4. Themagnetizing current I_(MAG) has a maximum magnitude defined by thefollowing equation:

$\begin{matrix}{I_{{MAG}\text{-}{MAX}} = {\frac{V_{BUS} \cdot T_{HC}}{4 \cdot L_{MAG}}.}} & \left( {{Equation}\mspace{14mu} 1} \right)\end{matrix}$where T_(HC) is the half-cycle period of the inverter voltage V_(INV),i.e., T_(HC)=T_(OP)/2. As shown in FIG. 4, the areas 250, 252 areapproximately equal, such that the average value of the magnitude of themagnetizing current I_(MAG) when the magnitude of the primary voltageV_(PRI) is greater than approximately zero volts.

The current sense circuit 260 averages the primary current I_(PRI)during the positive cycles of the inverter voltage V_(INV), i.e., whenthe high-side FET Q210 is conductive. The load current feedback signalV_(I-LOAD) generated by the current sense circuit 260 has a DC magnitudethat is the average value of the primary current I_(PRI) when thehigh-side FET Q210 is conductive. Because the average value of themagnitude of the magnetizing current I_(MAG) is approximately zero whenthe high-side FET Q210 is conductive, the load current feedback signalV_(I-LOAD) generated by the current sense circuit is representative ofonly the real component of the primary current I_(PRI).

The current sense circuit 260 comprises an averaging circuit forproducing the load current feedback signal V_(I-LOAD). The averagingcircuit may comprise a low-pass filter having a capacitor C230 (e.g.,having a capacitance of approximately 0.066 uF) and a resistor R232(e.g., having a resistance of approximately 3.32 kΩ). The low-passfilter receives the sense voltage V_(SENSE) via a resistor R234 (e.g.,having resistances of approximately 1 kΩ). The current sense circuit 160further comprises a transistor Q236 (e.g., a FET as shown in FIG. 2)coupled between the junction of the resistors R232, R234 and circuitcommon. The gate of the transistor Q236 is coupled to circuit commonthrough a resistor R238 (e.g., having a resistance of approximately 22kΩ) and receives the signal-chopper control signal V_(CHOP) from thecontroller.

When the high-side FET Q210 is rendered conductive, the controllerdrives the signal-chopper control signal V_(CHOP) low towards circuitcommon to render the transistor Q236 non-conductive for a signal-choppertime T_(CHOP), which is approximately equal to the on time T_(ON) of thehigh-side FET Q210 as shown in FIG. 4. The capacitor C230 is able tocharge from the sense voltage V_(SENSE) through the resistors R232, R234while the signal-chopper control signal V_(CHOP) is low, such that themagnitude of the load current feedback signal V_(I-LOAD) is the averagevalue of the primary current I_(PRI) and is thus representative of thereal component of the primary current during the time when the high-sideFET Q210 is conductive. When the high-side FET Q210 is not conductive,the controller 150 drives the signal-chopper control signal V_(CHOP)high to render the transistor Q236 non-conductive. Accordingly, thecontroller is able to accurately determine the average magnitude of theload current I_(LOAD) from the magnitude of the load current feedbacksignal V_(I-LOAD) since the effects of the magnetizing current I_(MAG)and the oscillations of the primary current I_(PRI) on the magnitude ofthe load current feedback signal V_(I-LOAD) are reduced or eliminatedcompletely.

As the target intensity L_(TRGT) of the LED light source 202 isdecreased towards the low-end intensity L_(LE) (and the on times T_(ON)of the drive control signals V_(DRIVE1), V_(DRIVE2) get smaller), theparasitic of the forward converter 140 (i.e., the parasitic capacitancesC_(P1), C_(P2) of the FETs, the parasitic capacitance C_(P3) of theprimary winding of the transformer 220, and other parasitic capacitancesof the circuit) can cause the magnitude of the primary voltage V_(PRI)to slowly decrease towards zero volts after the FETs Q210, Q212 arerendered non-conductive.

FIG. 5 shows example waveforms illustrating the operation of a forwardconverter and a current sense circuit (e.g., the forward converter 240and the current sense circuit 260) when the target intensity L_(TRGT) isnear the low-end intensity L_(LE). The gradual drop off in the magnitudeof the primary voltage V_(PRI) allows the primary winding to continue toconduct the primary current I_(PRI), such that the transformer 220continues to delivered power to the secondary winding after the FETsQ210, Q212 are rendered non-conductive as shown in FIG. 5. In addition,the magnetizing current I_(MAG) continues to increase in magnitude.Accordingly, the controller 150 increases the signal-chopper timeT_(CHOP) (during which the signal-chopper control signal V_(CHOP) islow) by an offset time T_(OS) when the target intensity L_(TRGT) of theLED light source 202 is near the low-end intensity L_(LE). Thecontroller may adjust the value of the offset time T_(OS) as a functionof the target intensity L_(TRGT) of the LED light source 202 as shown inFIG. 6. For example, the controller may adjust the value of the offsettime T_(OS) linearly with respect to the target intensity L_(TRGT) whenthe target intensity L_(TRGT) is below a threshold intensity L_(TH)(e.g., approximately 10%) as shown in FIG. 6.

FIG. 7 is a simplified flowchart of a control procedure 300 executedperiodically by a controller (e.g., the controller 150 of the LED driver100 shown in FIG. 1 and/or the controller controlling the forwardconverter 240 and the current sense circuit 260 shown in FIG. 2). Thecontroller may execute the control procedure 300, for example, at theoperating period T_(OP) of the inverter voltage V_(INV) of the forwardconverter 240. First, the controller determines the appropriate on timeT_(ON) for the drive control signals V_(DRIVE1), V_(DRIVE2) in responseto the target intensity L_(TRGT) and the load current feedback signalV_(I-LOAD) at step 310. If the controller should presently control thehigh-side FET Q210 at step 312, the controller drives the first drivecontrol signal V_(DRIVE1) high to approximately the supply voltageV_(CC) for the on time T_(ON) at step 314. If the target intensityL_(TRGT) is greater than or equal to the threshold intensity L_(TH) atstep 316, the controller 150 sets the signal-chopper time T_(CHOP) equalto the on time T_(ON) at step 318. If the target intensity L_(TRGT) isless than the threshold intensity L_(TH) at step 316, the controllerdetermines the offset time T_(OS) in response to the target intensityL_(TRGT) at step 320 (e.g., using the relationship shown in FIG. 6), andsets the signal-chopper time T_(CHOP) equal to the sum of the on timeT_(ON) and the offset time T_(OS) at step 322.

Next, the controller drives the signal-chopper control signal V_(CHOP)low towards circuit common for the signal-chopper time T_(CHOP) at step324. The controller then samples the averaged load current feedbacksignal V_(I-LOAD) at step 326 and calculates the magnitude of the loadcurrent I_(LOAD) using the sampled value at step 328, for example, usingthe following equation:

$\begin{matrix}{{I_{LOAD} = \frac{n_{TURNS} \cdot V_{I\text{-}{LOAD}} \cdot T_{HC}}{R_{SENSE} \cdot \left( {T_{CHOP} - T_{DELAY}} \right)}},} & \left( {{Equation}\mspace{14mu} 2} \right)\end{matrix}$where T_(DELAY) is the total delay time due to the turn-on time and theturn-off time of the FETs Q210, Q212, e.g.,T_(DELAY)=T_(TURN-ON)−T_(TURN-OFF), which may be equal to approximately200 μsec. Finally, the control procedure 300 exits after the magnitudeof the load current I_(LOAD) has been calculated. If the controllershould presently control the low-side FET Q210 at step 312, thecontroller drives the second drive control signal V_(DRIVE2) high toapproximately the supply voltage V_(CC) for the on time T_(ON) at step330, and the control procedure 300 exits without the controller drivingthe signal-chopper control signal V_(CHOP) low.

Alternatively, the controller can use a different relationship todetermine the offset time T_(OS) throughout the entire dimming range ofthe LED light source (i.e., from the low-end intensity L_(LE) to thehigh-end intensity L_(HE)) as shown in FIG. 8. For example, thecontroller could use the following equation:

$\begin{matrix}{{T_{OS} = \frac{\frac{V_{BUS}}{4} \cdot C_{PARASITIC}}{{\frac{T_{ON} + T_{{OS}\text{-}{PREV}}}{T_{HC}} \cdot I_{{MAG}\text{-}{MAX}}} + {\frac{K_{RIPPLE}}{n_{TURNS}} \cdot I_{LOAD}}}},} & \left( {{Equation}\mspace{14mu} 3} \right)\end{matrix}$where T_(OS-PREV) is the previous value of the offset time, K_(RIPPLE)is the dynamic ripple ratio of the output inductor current I_(L) (whichis a function of the load current I_(LOAD)) i.e.,K _(RIPPLE) =I _(L-PK) /I _(L-AVG),  (Equation 4)and C_(PARASITIC) is the total parasitic capacitance between thejunction of the FETs Q210, Q212 and circuit common.

As previously mentioned, the controller increases and decreases the ontimes T_(ON) of the drive control signals V_(DRIVE1), V_(DRIVE2) forcontrolling the FETs Q210, Q212 of the forward converter 140 torespectively increase and decrease the intensity of the LED lightsource. Due to hardware limitations, the controller may be operable toadjust the on times T_(ON) of the drive control signals V_(DRIVE1),V_(DRIVE2) by a minimum time step T_(STEP), which results in acorresponding step I_(STEP) in the load current I_(LOAD). Near thehigh-end intensity L_(HE), this step I_(STEP) in the load currentI_(LOAD) may be rather large (e.g., approximately 70 mA). Since it isdesirable to adjust the load current I_(LOAD) by smaller amounts, thecontroller is operable to “dither” the on times T_(ON) of the drivecontrol signals V_(DRIVE1), V_(DRIVE2), e.g., change the on timesbetween two values that result in the magnitude of the load currentbeing controlled to DC currents on either side of the target currentI_(TRGT).

FIG. 9 shows an example waveform of a load current conducted through anLED light source (e.g., the load current I_(LOAD) conducted through theLED light source 102). For example, the load current I_(LOAD) shown inFIG. 9 may be conducted through the LED light source when the targetcurrent I_(TRGT) is at a steady-state value of approximately 390 mA. Acontroller (e.g., the controller 150 of the LED driver 100 shown in FIG.1 and/or the controller controlling the forward converter 240 and thecurrent sense circuit 260 shown in FIG. 2) may control a forwardconverter (e.g., the forward converter 140, 240) to conduct the loadcurrent I_(LOAD) shown in FIG. 9 through the LED light source. Thecontroller adjusts the on times T_(ON) of the drive control signalsV_(DRIVE1), V_(DRIVE2) to control the magnitude of the load currentI_(LOAD) between two DC currents I_(L-1), I_(L-2) (e.g., approximately350 mA and 420 mA, respectively) that are separated by a step I_(STEP)(which is dependent upon the digital resolution of the LED driver 100and the impedance of the LED light source 102). In other words, thecontroller is operable to pulse-width modulate the magnitude of the loadcurrent I_(LOAD) between two DC currents I_(L-1), I_(L-2). The loadcurrent I_(LOAD) is characterized by a dithering frequency f_(DITHER)(e.g., approximately 2 kHz) and a dithering period T_(DITHER) as shownin FIG. 9. For example, a duty cycle DC_(DITHER) of the load currentI_(LOAD) may be approximately 57%, such that the average magnitude ofthe load current I_(LOAD) is approximately equal to the target currentI_(TRGT) (e.g., approximately 390 mA for the example of FIG. 9). Thedithering frequency f_(DITHER) is high enough that the human eye doesnot detect the change in the magnitude of the load current I_(LOAD)between the two DC currents I_(L-1), I_(L-2) when the target currentI_(TRGT) is at a steady-state value.

FIG. 10 shows an example waveform of the load current I_(LOAD) when thetarget current I_(TRGT) is dynamically changing (i.e., not at asteady-state value). For example, the target current I_(TRGT) is beingincreased with respect to time as shown in FIG. 10. The controller isable to adjust the on times T_(ON) of the drive control signalsV_(DRIVE1), V_(DRIVE2) to control the magnitude of the load currentI_(LOAD) between two DC currents I_(L-1), I_(L-2) that are separated bythe step I_(STEP). The duty cycle DC_(DITHER) of the load currentI_(LOAD) increases as the target current I_(TRGT) increases. At somepoint, the controller is able to control the on times T_(ON) of thedrive control signals V_(DRIVE1), V_(DRIVE2) to achieve the desiredtarget current I_(TRGT) without dithering the on times, thus resultingin a constant section 400 of the load current I_(LOAD). As the targetcurrent I_(TRGT) continues to increase after the constant section 400,the controller is able to control the on times T_(ON) of the drivecontrol signals V_(DRIVE1), V_(DRIVE2) to dither the magnitude of theload current I_(LOAD) between the DC current I_(L-2) and a larger DCcurrent I_(L-3).

However, the constant section 400 of the load current I_(LOAD) as shownin FIG. 10 may cause the human eye to detect a visible step in theadjustment of the intensity of the LED light source. Therefore, when thecontroller is actively adjusting the intensity of the LED light source,the controller is operable to control the magnitude of the load currentI_(LOAD) in response to the sum of the target current I_(TRGT) and asupplemental signal (e.g., a disturbance function). For example, thecontroller may be operable to control the average magnitude of the lampcurrent to the sum of the target current I_(TRGT) and the supplementalsignal. The supplemental signal may comprise, for example, a periodicsignal, such as a periodic ramp signal I_(RAMP) or sawtooth waveform asshown in FIG. 11. Note that the waveform of the ramp signal I_(RAMP)shown in FIG. 11 is a digital waveform, is not to scale, and is providedas an example. The ramp signal I_(RAMP) may be characterized by a rampfrequency f_(RAMP) (e.g., approximately 238 Hz) and a ramp periodT_(RAMP). When the controller is actively adjusting the intensity of theLED light source, a peak magnitude I_(RAMP-PK) of the ramp signalI_(RAMP) is set equal to a maximum ramp signal magnitude I_(RAMP-MAX)(e.g., approximately 150 mA). The ramp signal I_(RAMP) may increase withrespect to time in, for example, approximately 35 steps across thelength of the ramp period T_(RAMP). When the controller adds the rampsignal I_(RAMP) to the target current I_(TRGT) to control the on timesT_(ON) of the drive control signals V_(DRIVE1), V_(DRIVE2), theresulting load current I_(LOAD) has a varying magnitude. As a result,the constant section 400 of the load current I_(LOAD) is avoided and theperception to the human eye of the visible steps in the intensity of theLED light source as the controller is actively adjusting the intensityof the LED light source is reduced.

When the target current I_(TRGT) returns to a steady-state value, thecontroller may stop adding the ramp signal I_(RAMP) to the targetcurrent I_(TRGT). For example, the controller may decrease (e.g., fade)the peak magnitude I_(RAMP-PK) of the ramp signal I_(RAMP) from themaximum ramp signal magnitude I_(RAMP-MAX) to zero amps across a periodof time after the target current I_(TRGT) has reached a steady-statevalue. Alternatively, the controller may decrease the ramp periodT_(RAMP) to zero seconds across a period of time after the targetcurrent I_(TRGT) has reached a steady-state value.

While FIG. 11 shows the ramp signal I_(RAMP) (i.e., a sawtooth waveform)that is added to the target current I_(TRGT), other periodic waveformscould be used. For example, a ramp signal or sawtooth waveform having adecreasing amplitude could be used.

FIG. 12 is a flowchart of a load current adjustment procedure 500executed periodically by a controller (e.g., the controller 150 of theLED driver 100 shown in FIG. 1 and/or the controller controlling theforward converter 240 and the current sense circuit 260 shown in FIG.2). During the load current adjustment procedure 500, the controller isconfigured to control the magnitude of the load current I_(LOAD) towardsthe sum of the target current I_(TRGT) and a supplemental signal (e.g.,the ramp signal I_(RAMP) shown in FIG. 11). When the target currentI_(TRGT) is not at a steady-state value (i.e., the intensity of the LEDlight source is presently being adjusted) at step 510, the controllersets the peak magnitude I_(RAMP-PK) of the ramp signal I_(RAMP) equal tothe maximum ramp signal magnitude I_(RAMP-MAX) (e.g., approximately 150mA) at step 512. The controller then controls the magnitude of the loadcurrent I_(LOAD) towards the target current I_(TRGT) plus the rampsignal I_(RAMP) at step 514, and the load current adjustment procedure500 exits.

If the target current I_(TRGT) is at a steady-state value at step 510,but the peak magnitude I_(RAMP-PK) of the ramp signal I_(RAMP) is notequal to zero amps at step 516, the controller fades (e.g., decreases)the peak magnitude I_(RAMP-PK) from the maximum ramp signal magnitudeI_(RAMP-MAX) to zero amps across a period of time at step 518, andcontrols the magnitude of the load current I_(LOAD) to be equal to thetarget current I_(TRGT) plus the ramp signal I_(RAMP) at step 514,before the load current adjustment procedure 500 exits. Alternatively,the controller may decrease the ramp period T_(RAMP) to zero secondsacross a period of time at step 518. When the target current I_(TRGT) isat a steady-state value at step 510 and the peak magnitude I_(RAMP-PK)of the ramp signal I_(RAMP) is equal to zero amps at step 516, thecontroller controls the magnitude of the load current I_(LOAD) towardsthe target current I_(TRGT) at step 520, and the load current adjustmentprocedure 500 exits.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein, but only by the appended claims.

What is claimed is:
 1. A load control device for controlling the amountof power delivered to an electrical load, the load control devicecomprising: a load regulation circuit operable to conduct a load currentthrough the electrical load and to control the amount of power deliveredto the electrical load; and a controller coupled to the load regulationcircuit; wherein, when a target current of the load current is in asteady state, the controller is operable to pulse-width modulate themagnitude of the load current between a current less than the targetcurrent and a current greater than the target current to adjust anaverage magnitude of the load current to the target current; andwherein, when the target current is dynamically changing, the controlleris operable to pulse-width modulate the magnitude of the load currentbetween a sum of a supplemental signal and a current less than thetarget current and a sum of the supplemental signal and a currentgreater than the target current to adjust the average magnitude of theload current to the sum of the target current and the supplementalsignal.
 2. The load control device of claim 1, wherein the supplementalsignal comprises a periodic ramp signal.
 3. The load control device ofclaim 2, wherein the controller sets a peak magnitude of the periodicramp signal to a maximum ramp signal magnitude when the target currentis dynamically changing.
 4. The load control device of claim 3, whereinthe controller is operable to decrease the peak magnitude of theperiodic ramp signal from the maximum ramp signal to zero amps across aperiod of time after the target current reaches a steady-state value. 5.The load control device of claim 3, wherein the controller is operableto decrease a ramp period of the periodic ramp signal to zero secondsacross a period of time after the target current reaches a steady-statevalue.
 6. The load control device of claim 2, wherein the controllercomprises a microprocessor and the periodic ramp signal is a digitalsignal.
 7. The load control device of claim 1, wherein the loadregulation circuit comprises an LED drive circuit for an LED lightsource.
 8. The load control device of claim 7, wherein the LED drivecircuit comprises a forward converter.
 9. The load control device ofclaim 1, wherein the supplemental signal comprises a sawtooth waveform.10. The load control device of claim 1, wherein, when the target currentis in a steady state, the controller is operable to adjust a duty cycleof the pulse-width modulated load current to adjust the averagemagnitude of the load current between the current less than the targetcurrent and the current greater than the target current; and wherein,when the target current is dynamically changing, the controller isoperable to adjust the duty cycle of the pulse-width modulated loadcurrent to adjust the average magnitude of the load current between thesum of the supplemental signal and the current less than the targetcurrent and the sum of the supplemental signal and the current greaterthan the target current.
 11. A method for controlling the amount ofpower delivered to an electrical load, the method comprising: conductinga load current through the electrical load; adjusting, when a targetcurrent of the load current is in a steady state, an average magnitudeof the load current to the target current by pulse-width modulating themagnitude of the load current between a current less than the targetcurrent and a current greater than the target current; and adjusting,when the target current is dynamically changing, the average magnitudeof the load current to a sum of the target current and a supplementalsignal by pulse-width modulating the magnitude of the load currentbetween a sum of the supplemental signal and a current less than thetarget current and a sum of the supplemental signal and a currentgreater than the target current.
 12. The method of claim 11, wherein thesupplemental signal comprises a periodic ramp signal.
 13. The method ofclaim 12, further comprising: setting a peak magnitude of the periodicramp signal to a maximum ramp signal magnitude when the target currentis dynamically changing.
 14. The method of claim 13, further comprising:decreasing the peak magnitude of the periodic ramp signal from themaximum ramp signal to zero amps across a period of time after thetarget current reaches a steady-state value.
 15. The method of claim 13,further comprising: decreasing a ramp period of the periodic ramp signalto zero seconds across a period of time after the target current reachesa steady-state value.
 16. The method of claim 11, wherein thesupplemental signal comprises a sawtooth waveform.
 17. The method ofclaim 11, wherein the electrical load comprises an LED light source. 18.The method of claim 11, further comprising: adjusting, when a targetcurrent is in a steady state, a duty cycle of the pulse-width modulatedload current to adjust the average magnitude of the load current betweenthe current less than the target current and the current greater thanthe target current; and adjusting, when the target current isdynamically changing, the duty cycle of the pulse-width modulated loadcurrent to adjust the average magnitude of the load current between thesum of the supplemental signal and the current less than the targetcurrent and the sum of the supplemental signal and the current greaterthan the target current.